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Energy Reports 8 (2022) 1133–1140 www.elsevier.com/locate/egyr

2022 The 4th International Conference on Clean Energy and Electrical Systems (CEES 2022), 2–4 April, 2022, Tokyo, Japan

More electric aircraft challenges: A study on 270 V/90 V interleaved bidirectional DC–DC converter

Mustafa Tahira,∗, Shoaib Ahmed Khana, Tahir Khana, Muhammad Waseema,∗, Danish Khana, Andres Annukb

a College of Electrical Engineering, Zhejiang University, No. 38 Zheda Road, Xihu District, Hangzhou 310027, China b Chair of Energy Application Engineering, Institute of Forestry and Engineering, Estonian University of Life Sciences, 51006 Tartu, Estonia

Received 29 May 2022; accepted 25 June 2022 Available online 2 July 2022

Abstract

The concept of more electric aircraft (MEA) has gained increased importance in the aviation industry to make the aircraft nergy efficient and reduce carbon footprints. MEA facilitates transition towards clean energy and accomplishing the United ations’ net-zero emission goal by 2050. An efficient and reliable dc–dc converter is an indispensable part of the MEA that

nterfaces 270VDC and 28VDC bus bars in aircraft applications. Dual Active Bridge is a well-known and promising topology for his application; however, the problem of current and voltage ripples deteriorates the life of the battery and filtering components. n addition, the mentioned voltage levels demand high voltage gain, which makes design challenging. To reduce the voltage ain requirement along with ripple minimization, a two-stage bidirectional converter is proposed by combing a three-phase nterleaved buck-boost converter with Dual Active Bridge. The three-phase interleaved bidirectional converter is modelled and nalysed using the state-space averaging method, and a controller is established. Simulation results demonstrate the efficacy f the design. The research carried out in this paper can serve as a reference and offer future research directions using the roposed idea, which might assist other researchers in developing an efficient design, eventually facilitating the MEA and arbon footprint reduction. 2022 The Author(s). Published by Elsevier Ltd. This is an open access article under the CC BY-NC-ND license

http://creativecommons.org/licenses/by-nc-nd/4.0/).

eer-review under responsibility of the scientific committee of the 4th International Conference on Clean Energy and Electrical Systems, CEES, 2022.

eywords: More electric aircraft (MEA); Interleaved bidirectional converter (IBC); Electrification; Clean energy; Green energy

1. Introduction

Climate change has drastic effects on human life. The aerospace industry is responsible for 2% of total CO2 emission, and the percentage will increase to 3% by 2050 [1,2] since air passenger traffic is increasing at an average rate of 7% annually [3]. Advisory Council for Aeronautics Research has set a goal to reduce 50% in CO2 emission and 80% in NOx emission [3]. The salient challenges in the aerospace industry involve improving

∗ Corresponding authors. E-mail addresses: [email protected] (M. Tahir), [email protected] (M. Waseem).

ttps://doi.org/10.1016/j.egyr.2022.06.084 352-4847/© 2022 The Author(s). Published by Elsevier Ltd. This is an open access article under the CC BY-NC-ND license (http: /creativecommons.org/licenses/by-nc-nd/4.0/).

eer-review under responsibility of the scientific committee of the 4th International Conference on Clean Energy and Electrical Systems, EES, 2022.

M. Tahir, S. Ahmed Khan, T. Khan et al. Energy Reports 8 (2022) 1133–1140

t i p t

c m s m i t t t ( i L a i S m r

2

c a i B

emission, reducing fuel consumption and cost [4,5]. Hydraulic, pneumatic and mechanical power sources were used in aerospace applications [6]; nonetheless, such power sources were bulky with higher maintenance cost [7]. NASA completed a study that shows 10% reduction in empty aircraft weight and 9% reduction in fuel consumption can be achieved by using more electrical technologies in aircraft [8]. In [9], authors showed that replacing the pneumatic system with an electrical system in Boeing-787 has reduced 20% fuel consumption and CO2 emission compared o the conventional Boeing-767 aircraft. Therefore, the concept of MEA is trendy and electrical power demand n modern aircraft has increased. The Airbus 380 and the Boeing 787 have bigger electrical systems than their redecessors. In an aircraft electrical system, there are diverse types of loads that require a variety of input voltage o operate. As a result, there is a need for reliable and efficient power electronics equipment [7].

Fig. 1. Layout of aircraft distribution network.

In MEA, 270 V and 28 V are typical bus voltage levels [4]. As shown in Fig. 1 [10], a bidirectional dc–dc onverter is indispensable for power flow between two bus bars. The bidirectional dc–dc converter works in boost ode (28 V to 270 V) to support the startup of power generation and back up the critical load using a battery storage

ystem in case of generator failure. During regular aircraft operation, a bidirectional dc–dc converter works in buck ode (270 V to 28 V) to power up the low voltage loads along with battery charging. In technical literature, both

solated and non-isolated dc–dc converters can be found. In MEA, galvanic isolation is an essential requirement o prevent fault propagation between bus bars [4,11]. Therefore, researchers have devoted more efforts to isolated opologies and Dual Active Bridge (DAB) is one of the most attractive topologies. Currently, the systems are moving owards higher power density with better performance by using wideband gap devices [12], Zero Voltage Switching ZVS) and Zero Current Switching (ZCS) [13–15]. Research on the combination of different topologies has gained mportance as it enables increased power density design with improved efficiency. Researchers in [14] combined the LC and DAB converters to maximize the performance. This paper proposes a two-stage topology where the design nd modelling of a three-phase interleaved bidirectional converter are emphasized since many studies have already nvestigated the DAB. The content structure of the paper is as follows: The proposed idea is presented in Section 2. ection 3 explains the working modes of the interleaved bidirectional converter. Section 4 provides mathematical odelling and controller design. Discussion and analysis of simulation results are covered in Section 5. Concluding

emarks are given in Section 6.

. Proposed idea

The main drawback of most topologies in avionic applications is enormous current ripple stress in output/input apacitors [4]. In addition, for MEA applications, the voltage gain requirement is pretty high. To get an efficient nd reliable performance with minimum input/output current and voltage ripples, a combination of three-phase nterleaved buck-boost converter with DAB converter is proposed to produce the best possible dc–dc converter.

esides, the two-stage approach can streamline the design by easing voltage gain requirements. The first stage

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d d c

i

t T

s s i

g d

offers minimum ripples in current and voltage, while the second stage (DAB) is efficient owing to soft switching and provides galvanic isolation, a mandatory requirement in MEA.

In the proposed structure, an interleaved bidirectional converter (IBC) is connected between 270 V (high voltage c bus) and 90 V (Intermediate dc-link). Dual Active Bridge (DAB) is employed between 90 V (Intermediate c-link) and 28 V (Low voltage bus), as depicted in Fig. 2. This paper focuses on the interleaved bidirectional onverter (IBC) employed between 270 V and 90 V.

Fig. 2. Proposed two-stage bidirectional dc–dc converter.

3. Analysis of IBC modes

The proposed topology works by combing IBC with DAB to charge and discharge the battery. In the proposed topology, IBC is responsible for power flow between High Voltage (HV) dc-bus and intermediate dc link. The circuit diagram of IBC is shown in Fig. 4, where 1st phase (La, Sa1, Sa2, Da1, Da2), the second phase (Lb, Sb1, Sb2, Db1, Db2) and third phase (Lc, Sc1, Sc2, Dc1, Dc2) are in parallel. CHV is the HV dc bus capacitor and CLV is the intermediate dc-link capacitor. Besides, UHV and ULV are HV dc bus voltage and intermediate dc-link voltage, respectively.

3.1. Boost mode

Under boost mode, IBC takes part in battery discharging as a front-end converter. Lower parallel switches Sa2, Sb2, Sc2 and upper parallel diodes Da1, Db1, Dc1 take part in boost mode. The duty cycle in boost can be calculated using (1).

D = 1 − ULV

UH V (1)

Lower switches (Sa2, Sb2, Sc2) have the same duty cycle with a 120◦ phase shift. There are six modes of operations n continuous conduction mode (CCM) when duty cycle >0.6Ts, as shown in Fig. 3.

Fig. 3. Working waveforms of IBC in boost mode.

Modes 1, 3, 5: Operation during modes 1, 3 and 5 is the same. The switches Sa2, Sb2, Sc2 are turned on during hese modes, and the current in all inductors increases linearly. At this time while CHV provides energy to load. he equivalent electrical circuit for these modes can be seen in Fig. 4(a).

Mode 2: In the first mode, all inductors were being charged. In the second mode, Sb2 is turned off, and the tored energy in Lb gets transferred to the load through the Db1. Switches Sa2 and Sc2 remain turned on during the econd mode, and current across La and Lc keep increasing linearly. The equivalent electrical circuit for mode 2 is llustrated in Fig. 4(b).

Mode 4: In mode 4, Sa2 and Sb2 are turned on and Sc2 is turned off. The current across La and Lb increases radually and Lc transfers energy to the load through Dc1. The equivalent electrical circuit during this mode is

emonstrated in Fig. 4(c).

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i c

4

4

m c i i e T

o

Fig. 4. Equivalent electrical circuits of three-phase interleaved converter in boost mode.

Mode 6: Sb2 and Sc2 are turned on in the last mode, and Lb and Lc get into the charging state. Meanwhile, Sa2

s turned off. As a result, the stored energy in La gets transferred to the load through Da1. The equivalent electrical ircuit for mode 6 can be seen in Fig. 4(d).

. Modelling and control

.1. Modelling

Interleaved bidirectional converter (IBC) is a nonlinear system. To analyse static and dynamic performance, a athematical model is crucial. IBC has two modes (buck/boost). For that, boost mode state-space modelling is

arried out in this paper and that of buck mode follows the same methodology. There are six modes of operation n boost and their equivalent circuits are given in Section 3. The mathematical modelling in all modes of operation s carried out to get a state-space model. The converter is considered in CCM, and conduction losses are not ncompassed in modelling. To model the system, inductor currents and bus voltage are taken as state variables. he intermediate dc-link voltage is taken as the input variable as illustrated in (2).

x = [iLa(t), iLb(t), iLc(t), u H V (t), ]T u = uLV (t) (2)

During boost mode, lower active switches are controlled by duty cycles D1, D2 and D3. Applying KCL and KVL n Fig. 4(b), circuit behaviour is expressed in (3).

In mode 2 ((1 − D2) Ts):⎧⎪⎪⎪⎪⎪⎪⎪⎪⎪⎪⎨⎪⎪⎪⎪⎪⎪⎪⎪⎪⎪⎩

La diLa(t)

dt = uLV (t)

Lb diLb(t)

dt = uLV (t) − u H V (t)

Lc diLc(t)

dt = uLV (t)

CH V duH V (t)

= iLb(t) − u H V (t)

(3)

dt Rload

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m

i

w i

4

f c o P a G

Using (3), the state-space matrix for mode two is defined in (4).

{ . x = A1x + B1u

y = C1x A1 =

⎛⎜⎜⎜⎜⎜⎝ 0 0 0 0

0 0 0 −1/Lb

0 0 0 0

0 1/CH V 0 −1/(RloadCH V )

⎞⎟⎟⎟⎟⎟⎠ B1 =

⎛⎜⎜⎜⎜⎜⎝ 1/La

1/Lb

1/Lc

0

⎞⎟⎟⎟⎟⎟⎠ (4)

By circuit analysis of remaining equivalent circuits of boost mode in Section 3, state-space matrices for the other odes of operation are obtained and given in (5)–(7). In mode 4: ((1 − D3) Ts):

A2 =

⎛⎜⎜⎜⎜⎜⎝ 0 0 0 0

0 0 0 0

0 0 0 −1/Lc

0 0 1/CH V −1/(RloadCH V )

⎞⎟⎟⎟⎟⎟⎠ B2 =

⎛⎜⎜⎜⎜⎜⎝ 1/La

1/Lb

1/Lc

0

⎞⎟⎟⎟⎟⎟⎠ (5)

In mode 6 ((1 − D1) Ts):

A3 =

⎛⎜⎜⎜⎜⎜⎝ 0 0 0 −1/La

0 0 0 0

0 0 0 0

1/CH V 0 0 −1/(RloadCH V )

⎞⎟⎟⎟⎟⎟⎠ B3 =

⎛⎜⎜⎜⎜⎜⎝ 1/La

1/Lb

1/Lc

0

⎞⎟⎟⎟⎟⎟⎠ (6)

In mode 1, 3, and 5 ((D1 + D2 + D3 − 2) T s):

A4 =

⎛⎜⎜⎜⎜⎜⎝ 0 0 0 0

0 0 0 0

0 0 0 0

0 0 0 −1/(RloadCH V )

⎞⎟⎟⎟⎟⎟⎠ B4 =

⎛⎜⎜⎜⎜⎜⎝ 1/La

1/Lb

1/Lc

0

⎞⎟⎟⎟⎟⎟⎠ (7)

Using state-space averaging and superposition theorem, the transfer function for inductor current to duty cycle s expressed in (8).

G iLi d j (s) = îLi (s)

d̂ j (s)

⏐⏐⏐⏐⏐ ûc(s)=0

=

UH V CH V s + (1 − D j )ILi + UH V Rload

L i CH V s2 + L i

Rload s + (1 − D j )2

(8)

here i = a, b, c. and j = 1, 2, 3. Correspondingly, the transfer function voltage to inductor current is expressed n (9).

Gu H V iLi (s) = ûH V (s)

îLi (s)

⏐⏐⏐⏐⏐ ûLV (s)=0

=

−ILi L i CH V s2 +

[ (1 − D j )UH V CH V −

ILi L i Rload

] s +

(1−D j )UH V Rload

(CH V s + 1

Rload )[UH V CH V s + (1 − D j )ILi +

UH V Rload

] (9)

.2. Controller

For both (boost & buck) modes, dual-loop control is implemented. The block diagram of the control system or boost mode with inner current loop and outer voltage loop is illustrated in Fig. 5. Three independent current ontrollers are employed to realize equal current sharing among interleaved inductors which is crucial for reliable peration. Duty cycles are generated by current loops and reference current is derived from sampled dc bus voltage. I controller is used because of its excellent performance, feasibility and ease of implementation. GPI−La, GPI−Lb nd GPI−Lc are current controller transfer functions. Likewise, GPI−U is voltage controller transfer function and

(s), G (s) and G (s) are modulation transfer functions. Carrier phase-shifted SPWM is employed to reduce

M1 M2 M3

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Table 1. Design parameters of the interleaved converter.

Parameter Symbol Value

Power P 6.6 kW HV side voltage UHV 270 V DC-link voltage ULV 90 V Inductor La Lb Lc 200 µH Frequency fs 100 kHz DC link filter capacitance CLV 30 µF DC bus filter capacitance CHV 150 µF

Fig. 5. Dual loop control block scheme for three-phase interleaved converter.

the current ripple. Using (8), (9) and parameters in Table 1, transfer functions of the system can be obtained. After adding a controller into the system, the crossover frequency is much smaller than the switching frequency. Frequency domain analysis of the system is completed using the bode plot. The phase margin value is set at 82 degree, and the PI controller is tuned to neutralize the small-signal disturbance for voltage and current controllers. The controller gain values are the same for all three current controllers. In boost mode, Uref is set equal to dc bus voltage and open-loop system analysis is completed in the frequency domain for the voltage controller. Phase and gain margins are observed. Considering the margins, the gain values for the controllers are tuned to achieve optimal performance.

5. Simulation results

To verify the efficacy of system, three phase IBC is simulated in MATLAB with a dual loop control at rated power of 6.6 kW. In boost mode, dc-link voltage acts as input voltage (90 V), beside bus voltage (270 V) is the output voltage and vice versa. Ripple compassion of the three-phase IBC converter and its steady-state and dynamic performance is validated through the simulation results.

Ripple reduction in the output voltage of three-phase IBC compared with single phase counterpart during the boost and buck mode is shown in Fig. 6(a–b) and (c), respectively. In addition, the output current ripple reduction in the buck mode is illustrated in Fig. 6(d). Likewise, Fig. 6(e) shows that the input current ripple of IBC in the boost mode is minimal. In addition, the total input current is the sum of inductor currents and equal current distribution (ILa = ILb = ILc) among inductors is achieved, which is crucial for converter reliability. Hence, results show that IBC has a much smaller ripple than its conventional counterpart.

To investigate the dynamic performance, load is increased from 3 kW to 6 kW and decreased back to 3 kW at 0.4 s and 0.8 s, respectively, for both buck/boost modes. For boost mode, it can be seen in Fig. 6(f) that bus voltage stabilized rapidly and overshoot is restrained within 3%. Moreover, all inductor currents trace the load mutation effectively, as depicted in Fig. 6(g). A similar response of current and voltage curves for buck mode is shown in Fig. 6(h). A drop in the input voltage (90 V to 80 V) is also investigated in boost mode. Fig. 6(i) reveals that the controller quickly stabilizes dc bus voltage during input voltage mutation, which shows that the designed controller provides the desired performance. Hence, results validate the performance of the designed converter for both modes of operations.

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Fig. 6. Simulation results.

6. Conclusion

An efficient and reliable bidirectional dc–dc converter is an indispensable part of More electric aircraft that is energy efficient and playing a curial role in meeting environmental goals. This paper proposes a two-stage bidirectional converter, which can infuse flexibility in topology, easing voltage gain and outrageous ripple reduction. An in-depth analysis of three-phase interleaved bidirectional converter is carried out. Firstly, working modes of BIC are analysed in detail, and then by using the state-space averaging method, modelling is completed, and expressions of transfer functions are derived. Steady-state and dynamic analysis is carried out using open-loop transfer functions, and a dual loop controller is established. Besides, current balancing among inductor currents is also considered as it is crucial for reliable converter operation. The authenticity of the proposed design is verified by simulation, which shows that the designed converter has much smaller ripples in output/input current and voltage. This study encompasses the in-depth analysis of the IBC stage, and combined analysis of both stages is a part of future work. In addition, the potential of the two-stage idea can be unleashed by exploiting the wideband gap devices that enable high efficiency and high power density design. The advanced control techniques can also be investigated on the proposed two-stage converter to unfold its benefits. Furthermore, the analysed IBC is suitable for a modular design approach to extend power ratings. The research carried out in this paper can serve as a reference for designing, modelling, and controlling dc–dc converter for other applications requiring low ripples.

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Declaration of competing interest

The authors declare that they have no known competing financial interests or personal relationships that could ave appeared to influence the work reported in this paper.

cknowledgements

The authors extend their appreciation to the Estonian Centre of Excellence in Zero Energy and Resource Efficient mart Buildings and Districts, ZEBE, grant TK146, funded by the European Regional Development Fund to support

his research.

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