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48 V-to-1 V Transformerless Stacked Active Bridge Converters with Merged Regulation Stage

Jianglin Zhu, and Dragan Maksimovic Colorado Power Electronics Center

Department of Electrical, Computer and Energy Engineering University of Colorado, Boulder, Colorado 80309

Email: [email protected]

ABSTRACT

Transformerless stacked active bridge (TSAB) convert- ers are hybrid converters derived from switched capacitor converters by addition of small inductors. TSAB converters achieve the highest efficiency around a nominal conversion ratio because of “soft” charging/discharging of all flying capacitors, low rms currents and zero-voltage-switching of power switches. This paper introduces new configurations of TSAB converters with inductive filter as opposed to ca- pacitive filter at the output port, which opens opportunities to merge a regulation stage at the output. The approach is applied to 48 V to 1 V point-of-load (PoL) conversion by merging a 6-to-1 Dickson TSAB with a multi-phase buck converter at the regulation stage, greatly reduced the need for a bulky intermediate bus capacitor. Experimental results are provided for a 48 V-to-1 V, 100 A prototype consisting of a 6-to-1 TSAB operating at 100-125 kHz using 120 nH inductors, and an off-the-shelf four-phase buck regulating stage operating at 500 kHz. The prototype has 91.5% peak efficiency at 25 A and greater than 85% efficiency up to 90 A.

I. INTRODUCTION

Motivated by the reduction in distribution losses, there is an increased interest in higher voltage, e.g. Vbus = 48 V, dc dis- tribution in data center applications, which in turn highlights the need for high step-down point-of-load (POL) conversion, e.g. 48-to-1 V, power xPU’s (GPUs, CPUs, and TPUs) on server boards. In a conventional design, a two-stage ap- proach is adopted, with an intermediate bus voltage VIB (e.g. VIB = 12 V), where the output stage is typically implemented using readily available multi-phase buck regulators. Various approaches have been pursued to improve the performance of the front-end Vbus-to-VIB stage, including switched-tank converters [1], [2], LLC converters [3], resonant SC converter [4], and TSAB converters [5]. To further improve the power density and efficiency performance, direct Vbus = 48 V to PoL converter topologies have also been investigated, such as the Sigma converter [6], and the DIHC hybrid converter [7], [8].

Vbus

vIB

L4C4

L2C2

Vout

c2s

c1s

iIB

vQ9

vQ1

Lb1

Lbn

Vbus

vIB

L4C4

L2C2

Vout

c2s

c1s

iIB

vQ9

vQ1

Lb1

Lbn

(b)

(a)

Q2Q3

Q10

Q2Q3

Q10

Fig. 1: Inductive-output 6-to-1 TSAB merged with a multi-phase buck regulation stage: (a) with a small LC filter in between, and (b) with no filter in between.

In transformer-isolated converters, core losses and con- duction losses in the transformer can be significant at high switching frequency and high turns ratio. Advanced techniques such as matrix transformer [9], and “integration” of secondary winding and rectifier [3] have been proposed to address such challenges. Since galvanic isolation is not required, trans- formerless solutions, such as hybrid converters, are potentially advantageous as they offer additional benefits such as reduced switch voltage stresses, as well as soft switching.

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VIN

L4C4

L2C2

voutLdc

VIN

vout

(a) (b)

c2s

c1s

Fig. 2: Two 6:1 Dickson TSAB converters: (a) capacitive-output TSAB where C2, C4 and the output filter capacitor are considered tree branches, and (b) inductive-output TSAB where the tree branches are C1, C3, C5. The tree branches are highlighted in bold in the two configurations.

converter with efficiency optimized at a nominal step-down conversion ratio. A step further has recently been proposed in [10], [11], where the operation of a front-end switched- capacitor stage is “merged” with a follow-up buck regulation stage. Compared with a decoupled two-stage solution, the front-end and the back-end converters are coupled in oper- ation so that filter components such as the intermediate bus capacitors can be reduced or eliminated. A similar approach is proposed in this paper based on merging a transformerless stacked active bridge (TSAB) converter and a multi-phase buck regulator.

TSAB converters [12] are non-isolated hybrid dc-dc con- verters with characteristics similar to the isolated Dual-Active- Bridge (DAB) converter [13]. TSAB converters can be de- rived from switched-capacitor (SC) converters by addition of small ac inductors to eliminate “hard” capacitor charg- ing/discharging losses [12]. The operation and control are similar to the DAB converter [14]: output can be continuously regulated by simple phase shift control, inductor peak current are close to minimum and ZVS operation can be achieved for most of the switches. Compared to other types of hybrid converters, such as resonant switched-capacitor converters [4], [15], or switched tank converters [2], [16], [17], which operate near resonance (fs/fr ≈ 1), TSAB converters are designed to operate above resonance (fs/fr > 1) with trapezoidal near- minimum RMS inductor currents, and zero-voltage-switching of power devices [12]. Above 98.5% efficiency and flat ef- ficiency curves have been experimentally demonstrated on a 4-to-1 Dickson-based TSAB [5], and a 3-to-1 ladder-based TSAB converter [18].

Two variants of a merged TSAB/regulation-stage converter proposed in this paper are shown Fig. 1. The front-end stage, which is a 6-to-1 Dickson TSAB with an inductive output filter, is merged with a standard multi-phase buck POL regulator. The variant in Fig. 1(a) retains a very small LC filter between the two stages, while the intermediate-bus filter is completely eliminated in the variant shown in Fig. 1 (b).

The paper is organized as follows: steady-state operation and soft switching TSAB converters with inductive output port are described in Section II. An analysis of the merged configu- rations shown in Fig. 1 is presented in Section III. Design of a 48 V-to-PoL, 100 A prototype is described in Section IV, along with key experimental results, including operating waveforms and efficiency curves. The prototype consisting of a 6-to-1 inductive-output Dickson TSAB operating at 100-125 kHz using 120 nH ac inductors, and an off-the-shelf four-phase buck regulating stage operating at 500 kHz has 91.5% peak efficiency at 25 A and greater than 85% efficiency up to 90 A. Section V concludes the paper.

II. INDUCTIVE-OUTPUT TSAB CONVERTERS

TSAB converters introduced in [5], [12], [18] employ one or more ac inductors to ensure soft charging and discharging of flying capacitors. The output port has a dc filter capacitor. The ac inductors and phase-shift operation yield trapezoidal current waveforms with low RMS currents, as well as zero voltage switching. As an example, the 6:1 Dickson TSAB converter is shown in Fig. 2(a). In this converter, each inductor shares the output current equally, and switches are operated with reduced voltage stress (Vout and 2Vout, respectively) and ZVS is achieved for most of the switches at sufficiently high load current [5]. In general, capacitive-output TSAB converters are obtained by inserting ac inductors into the link branches of the corresponding two-phase switched-capacitor converter. If the links remain the same in each switched state, hard charging/discharging loops consisting of capacitors only are eliminated.

As discussed in [12], the designation of links and tree branches in an SC converter is not necessarily unique. In the 6:1 Dickson SC converter, if the output branch Vout is treated as a tree branch, the tree capacitors are C2 and C4 and the link capacitors are C1, C3, and C5. Inserting ac inductors in series with the link capacitors yields the 6-to-1 Dickson TSAB shown in Fig. 2(a). If, however, the output branch Vout is treated as a link, the link capacitors are C2 and C4, while the tree capacitors are C1, C3, and C5. Two ac inductors, L2 and L4 are inserted in series with C2 and C4, respectively, while a dc filter inductor Ldc is in the output branch, as shown in Fig. 2(b). It should be noted that the dc filter inductor Ldc carries the dc output current and ideally does not withstand any volt-seconds , so that the inductance can be very small, and inductor losses can be negligible compared to the ac inductors which carry large ac currents.

A. Steady-state operation

In the inductive-output TSAB of Fig. 2(b), there are four switched states, as illustrated in Fig. 4. States 1 and 2 are the power-delivering states, where the inductors withstand zero volt-seconds; states 1’ and 2’ are the polarity-reversal states where the ac inductor currents are flipping polarities. Assuming operation at nominal output voltage, which is the same as in the original SC converter, the dc filter inductor does not withstand any volt-seconds, so a very small inductance can

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c1s

c2s

iL2 iL4

IIB/3

iLdc IIB

Tφ

j 1 2 11'2' 2 1'2'

Fig. 3: Ideal operating waveforms in the inductive-output 6:1 Dickson TSAB converter in Fig. 2(b). c1/c2, c1s/c2s are complementary 50% signals with phase shift Tϕ. Ideal operation assumes nominal operation above resonance fs � fr. Dead times are ignored, which is why there is no ripple in the output filter inductor current iLdc.

(a) State 1'

VIN

C5 VOUT Ldc

L2C2

C3 L4C4

C3 L2

C2 C1

VOUT Ldc

VIN

C5 VOUT Ldc

L2 C2

VOUT Ldc

L4 C4

(b) State 2'

vx vx

vxvx

Fig. 4: Four switched-states in the inductive-output TSAB of Fig. 2(b). States 1 and 2 are power-delivering states, and states 1’ and 2’ are polarity-reversal states.

be employed. Trapezoidal ac inductor currents can be realized by introducing phase shifts between the control signals of the two legs of the full-bridge rectifier (Q1 − Q4). Four control signals c1,c2 and c1s, c2s are required, and the corresponding timing diagram is shown in Fig. 3.

The balanced flying capacitor voltages are:

Vout ≈ VC1 ≈ Vbus − VC5 (1)

By symmetry, VC3 ≈ Vbus/2 (2)

By volt-seconds balance applied to L2 and L4,

VC2 = VC1 + VC3

2 (3)

VIN

L4C4

L2C2

c2s

c1s Q1

Q2 D3

Q10

vout

VIN

L4C4

L2C2

c2s

c1s Q1

Q2 D3 D4

Q10

voutLdcLdc

(a) (b)

Fig. 5: Zero voltage switching transitions during (a) dead-time before c2 turns on, and (b) dead-time before c1 turns on.

VC4 = VC3 + VC5

2 (4)

Similar to the capacitive-output TSAB converter shown in Fig. 2(a), the average output current is controlled by the phase shift:

ILdc = Vbus

8Lacfs ϕ(1 − ϕ) (5)

where ϕ = 2Tϕ/Ts, and Lac = L2 = L4. Similar to a DAB converter, operating away from the nominal voltage results in increased volt-seconds on inductors and consequently in- creased RMS currents.

B. Zero voltage switching transitions

For the full-bridge switches Q1 − Q4, inductor current of Ldc forward biases the body diodes of either Q1,Q2 or Q3,Q4 during dead-times, so zero voltage switching can always be achieved. Fig. 5 illustrates a ZVS transition for top switches Q6 − Q9: the body diodes of these switches are forward- biased by the inductor currents during the dead-time, assuming sufficiently large inductive energy storage, i.e., sufficiently large load current. Similar analysis shows that for the left of the switches Q5,Q10, ZVS cannot be achieved. Importantly, the ZVS switches Q6 − Q9 also block the highest voltage (2Vout), while the hard switched devices only block Vout.

III. INDUCTIVE-OUTPUT TSAB CONVERTER MERGED WITH A REGULATION STAGE

This section describes how an inductive-output TSAB con- verter can be used as a front-end stage in a high step-down application such as 48 V-to-PoL conversion. The follow-up PoL converter can be a standard multi-phase buck converter, as shown in Fig. 1(a). Furthermore, the small intermediate-bus LC filter can be completely eliminated as shown in Fig. 2(b). In this variant, the buck inductor serve the function of the output inductor Ldc.

Eliminating the LC intermediate-bus filter imposes two constraints. First, higher RMS currents are induced in some of the TSAB capacitors and switches. The switching frequency of the buck converter is typically much higher than the switching

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frequency of the TSAB converter. As a result, the buck induc- tor current flows through C1 and C5 alternatively in states 1 and 2, which increases the RMS currents in capacitors C1 and C5 and the TSAB corresponding switches. The inductor currents iL2 ad iL4 however, remain essentially unaffected.

Additionally, to ensure balanced voltages in the TSAB converter, switching frequency ratio between the TSAB and the buck stages should be an even integer value in the converter of Fig. 1(b). If a multi-phase buck stage is employed, as shown in Fig. 1(b), interleaving yields an equivalent switching frequency proportional to the number of phases. The RMS currents supplied by C1 and C5 can then be much smaller.

IV. DESIGN AND EXPERIMENTAL RESULTS FOR A 48 V-TO-1 V PROTOTYPE

This section presents the design and the experimental results for a 48V-to-1 V prototype based on the merged TSAB/regulation-stage converter shown in Fig. 1.

A. 48V-to-PoL prototype design

The design process for the 6-to-1 TSAB in Fig. 1 is summarized in this section.

LTM4680

L2 L4

LTM4680

Fig. 6: 48 V-to-PoL/100 A prototype including 6:1 inductive-output TSAB with planar inductors followed by a four-phase buck regulator. L2,L4 are plannar inductors with 4 mm in height.

1) Selection of TSAB L and C values: The choice of the inductive impedance ZL = ωsL and capacitive impedance ZC = 1/(ωsC) affects the TSAB RMS currents. Let Rp be the series resistance in the tanks consisting of ac inductors L2, L4 and link capacitors C2, C4, respectively. Assuming constant Rp, the quality factor Q =

√ (ZLZC)/Rp and the

ratio k = fs/fr = √ ZL/ZC determine the waveshapes of the

ac inductor currents. As can be seen from simulations, a high Q (Q > 10) and low k (k < 1.5) result in near-sinusoidal current waveforms, whereas a low Q (Q < 3) and high k (k > 1.5) result in near-trapezoidal waveforms. The k and Q values also affect the required value of the phase shift for a given output current. A larger phase shift corresponds to a larger circulating current.

Switches

Q1 − Q4,Q5, Q10 EPC2023

Q6 − Q9 EPC2020

Passive components

C1 48 µF

C3,C5 80 µF

C2,C4 72 µF

L2,L4 120 nH

fsw,tsab 100-125 kHz

fsw,buck 500 kHz

k 1.7

Q ≈ 6

Inductor design

Number of turns 1

Core PC95 ELT11x4

AC resistance 4 mΩ

TABLE I: Parameters in the 48-to-1 V, 100 A prototype

In the prototype design, k = 1.7, Q = 10 and fs = 100 kHz are selected, which ensures quasi-trapezoidal ac inductor cur- rents with less than 2% required phase shifts at full load. The corresponding parameter values are listed in Table I.

2) Design of ac inductors: Two low-profile 120 nH ac inductors are implemented as planar low-profile inductors (4 mm in height). The core material is PC95. Low inductances allows for a single-turn design, which eliminates ac copper losses due to proximity effects. Since the air gap is on the opposite side of the PCB winding, the losses due to fringing field are also relatively small. At the switching frequency, the ac resistance obtained by finite-element simulation is found to be very close to the dc resistance. Inductor core losses are relatively low compared to the copper losses due to the low peak-to-peak flux density.

3) Loss modeling: Major losses in the TSAB stage include switch losses (conduction, switching losses), magnetic losses (ac copper loss and core loss), and capacitor ESR losses.

In this design with four phase interleaved buck follow-up stage, the RMS currents in the TSAB switches depend on the nature of the intermediate-bus filter. If a large LC filter with cutoff frequency lower than twice the TSAB switching frequency is used, the RMS currents are minimized. On the other hand, if a small (or no) filter is used, the RMS currents are increased. In this case, the conduction loss model must take into account the input current ripple in the follow-up buck regulation stage.

Regarding switching losses, note that ZVS is achieved for most of the TSAB switches assuming sufficient load current. For the hard switching devices, and for the soft switching devices at light load, switching losses are estimated by taking into account voltage/current overlap losses and Coss losses.

Inductor copper losses are estimated based on ac resistance found by finite-element analysis as shown in Table I, while core losses are estimated using the iGSE method [19],.

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4) Multi-phase buck converter: The multi-phase buck con- verter used in the prototype is based on two off-the-shelf LTM4680 components in parallel [20], to obtain a four-phase configuration with current mode control.

B. Experimental results

A 48 V-to-1 V converter prototype capable of 100 A output current is shown in Fig. 6, and the component values are listed in Table I.

Switch node voltages and ac inductor currents in the TSAB stage in the configuration shown in Fig. 1 (a) are shown in Fig. 7. Dips in vIB in Fig. 7 is attributed to dead-time between signal c1/c2 and c1s/c2s. For example. during the dead-time of Q1/Q2, both switch operate in reverse conduction mode, and vIB collapses.

vIB(10V/div)

vQ9(12.5V/div)

iL2(5A/div)

iL4(4A/div)

Fig. 7: Waveforms in the TSAB stage with an LC filter: Vbus = 48 V, VIB = 7.84 V, IIB = 15 A, fs = 100 kHz, time division: 4 µs/div. vQ9 and vIB are as shown in Fig. 1(a).

Efficiencies are measured separately for the TSAB and for the four-phase buck stage from 10 A to 90 A load current for Vbus = 48 V, Vout = 1 V as shown in Fig. 8. The TSAB efficiency remains above 97.5% across the measured load range with peak efficiency of 98.2% at 40 A. The four- phase buck stage is operated with the four phases interleaved, and the peak efficiency is around 94% at 20 A. The resultant system peak efficiency is 91.5% at 25 A, and drops to 85% at 90 A. A small filter inductor Lf ≈ 60 nH and filter capacitors Cf ≈ 22µF is used for each buck module. The peak efficiency is comparable with other 48 V-to-1 V hybrid converter solutions, such as DIHC [21], and heavy- load efficiencies are improved. The heavy-load efficiency is comparable to or slightly higher compared to transformer- based two stage solutions, such as the current doubler design reported in [22]. Additionally, compared with other direct 48 V-to-1 V solutions, regulation is much simpler because readily available off-the-shelf buck modules can be employed. Estimated loss breakdown in the TSAB stage for 1 V/40 A output is shown in Fig. 9. The load current of the TSAB stage is around 5 A, with longer than minimum dead times required to achieve ZVS. In the model, the switch conduction losses includes Rds,on losses as well as losses due to the PCB trace resistances, and reverse-conduction losses are related to the voltage drops across GaN FETs during dead-times.

As explained in Section III, removing the LC filter in Fig. 1(b) is advantageous for size reduction. Fig. 10 compares measured efficiency in the no-filter configuration of Fig. 1(b)

20 40 60 80 100 Output Current [A]

100

Buck

System

TSAB

91.5%

E ff ic

ie n cy

[ %

]

Fig. 8: Measured efficiencies of the 6:1 TSAB stage and the 8:1 four-phase buck converter. The system efficiency is shown for the configuration in Fig. 1(a) at 48 V input and 1 V output.

and small LC filter configuration of Fig. 1 (a). The small LC filter yields improvement in efficiency in this case because the RMS current in the switches are reduced compared to the case with no filter. In terms of operation, LC filter is not necessary. The operating waveforms for no filters are shown in Fig. 11. Even with no input capacitors, the buck input voltage (vIB) ripple is less than 1 V. This is because the flying capacitors, especially C1 and C5 serve as an effective filter for the input current ripple.

Conduction losses 27%

Switching losses 35%

Reverse-conduction losses 28%

Lac losses 8%

Ldc losses 1%

ESR losses 1%

Fig. 9: Modeled loss breakdown for the 6:1 TSAB prototype with Vbus = 48 V input, 1 V output, at the output current of 40 A.

V. CONCLUSIONS

TSAB converters [5], [12], [18] employ small ac induc- tors while operating at relatively low switching frequency, and offer high efficiency at nominal conversion ratio due to “soft” charging/discharging of all capacitors and zero-voltage- switching of power devices. TSAB converters can be cascaded with a standard multi-phase buck regulation stage to construct 48 V-to point-of-load voltage around 1 V for server and

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Fig. 10: Measured efficiency for the configuration with a small LC filter (Lf =60 nH, Cf = 8µF ) in Fig. 1 (a) and the no-filter configuration in Fig. 1(b). Operating conditions: Vbus = 40 V, Vout = 1 V, fs,T SAB = 125 kHz, fs,buck = 500 kHz.

vQ9(12.5V/div)

Buck switch node (5V/div)

vIB(1V/div)

iL4(2A/div)

Fig. 11: Experimental waveforms in the no-filter configuration in Fig. 1(b): Vbus=30 V, Vout = 1 V, Iout = 30 A, fs,T SAB = 125 kHz, fs,buck = 500 kHz. Time division: 2 µs/div. vQ9 and vIB are as shown in Fig. 1(b).

other applications. To further reduce the losses associated with ac inductors, this paper presents a new configuration of the TSAB converter with an inductive output which eliminates one of ac inductors. A follow-up multi-phase buck regulation stage can be merged with the inductive-output TSAB using a small LC filter. Completely removing the intermediate filter is also feasible with a slight trade-off in conduction losses. An experimental prototype is constructed using a 6-to-1 inductive- output Dickson TSAB operating at 100-125 kHz followed by an off-the-shelf four-phase buck regulator operating at 500 kHz. The prototype has 91.5% peak efficiency at 25 A and greater than 85% efficiency up to 90 A.

ACKNOWLEDGMENT The authors would like to acknowledge Lockheed Martin

for supporting research reported in this paper.

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